2016 Circular Polarizaration reflecct arrays.pdf

2016 Circular Polarizaration reflecct arrays.pdf

1. IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 10, OCTOBER 2016 4235
Broadband, Single-Layer Dual Circularly Polarized
Reflectarrays With Linearly Polarized Feed
Geng-Bo Wu, Shi-Wei Qu,Senior Member, IEEE, Shiwen Yang,Senior Member, IEEE,
and Chi Hou Chan,Fellow, IEEE
Abstract— A broadband single layer, dual circularly
polarized (CP) reflectarrays with linearly polarized feed is
introduced in this paper. To reduce the electrical interference
between the two orthogonal polarizations of the CP element,
a novel subwavelength multiresonance element with a Jerusalem
cross and an open loop is proposed, which presents a broader
bandwidth and phase range excessing 360° simultaneously.
By tuning thex-andy-axis dimensions of the proposed element,
an optimization technique is used to minimize the phase errors
on both orthogonal components. Then, a single-layer offset-fed
20×20-element dual-CP reflectarray has been designed and
fabricated. The measured results show that the 1-dB gain and
3-dB axial ratio (AR) bandwidths of the dual-CP reflectarray can
reach 12.5% and 50%, respectively, which shows a significant
improvement in gain and AR bandwidths as compared to
reflectarrays with conventionalλ/2 cross-dipole elements.
Index Terms— Broadband, circularly polarization (CP),
reflectarray, single layer.
I. INTRODUCTION
M
ICROSTRIP reflectarrays are rapidly becoming an
attractive alternative to the traditional parabolic
reflectors for wireless communications due to their various
advantages, e.g., low profile,low cost and simplification of
polarization transformation, etc., [1], [2]. In satellite commu-
nications, the robustness against environmental interference
of the circularly polarized (CP) antenna is considered to be
of significant importance due to the Faraday rotation effect
of the ionized layer of the atmosphere. Therefore, a variety
of different CP reflectarrays have been proposed and imple-
mented [3]–[9]. In the past decades, two different methods
have been proposed to design a CP reflectarray.
In the first method, reflectarrays can achieve circular
polarization using a CP feed, and their elements can be
Manuscript received February 6, 2016; revised June 28, 2016; accepted
July 19, 2016. Date of publication July 22, 2016; date of current version
October 4, 2016. This work was supported in part by the Sichuan Youth
Science and Technology Foundationunder Grant 2014JQ0012, in part by the
Natural Science Foundation of China under Project 61371051, and in part by
the Research Grants Council of the Hong Kong SAR, China, under Project
CityU110713.
G.-B. Wu, S.-W. Qu, and S. Yang are with the School of Elec-
tronic Engineering, University of Electronic Science and Technol-
ogy of China, Chengdu 611731, China (e-mail: [email protected];
[email protected]; [email protected]).
C. H. Chan is with the State Key Laboratory of Millimeter Waves,
Partner Laboratory, City University of Hong Kong, Hong Kong (e-mail:
[email protected]).
Color versions of one or more of the figures in this paper are available
online at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2016.2593873
variable-size patch elements, variable-length delay elements,
or angular rotation elements [3]–[6], etc. However, a CP feed
antenna as a primary source of the reflectarray is necessary
in this method, which introduces more complexity in design
and fabrication of the CP feed, especially in millimeter wave
or higher frequency bands.
The second method to design a CP reflectarray is to use
a linear polarized (LP) feed. The reflectarray elements are
designed to transform LP incident waves into CP reflected
waves and to collimate the beam simultaneously. Many kinds
of elements can be used to design the reflectarray with the
LP feed, e.g., single-layer rectangular patch elements [7],
dual-layer T-shape elements [8], or aperture coupled patch
elements with slots and lines of variable lengths [9], etc.
However, both the phase linearity and the phase dynamic
range of the CP element limit the bandwidth of this kind
of CP reflectarray [10]. Moreover, the mutual coupling
between the two orthogonal field components of CP reflec-
tarray element is usually neglected to make separated phase
tuning for both linear polarizations, leaving the control of
cross coupling alone. For example, a single-layer subwave-
length rectangular element in [7] has only one resonance
and its inadequate phase range results in a small 3-dB
axial ratio (AR) bandwidth of the reflectarray. Meanwhile, the
cross coupling between orthogonal field components of the
rectangular element is relatively high which will be explained
in Section II of this paper. As another example, a dual-layer
T-shape element structures in [8] have been employed to
provide an adequate phase range over 360° and the 1-dB gain
bandwidth and 3-dB AR bandwidth of the reflectarray reach
20% and 28%, respectively. However, the employed multilayer
configuration leads to additional fabrication complexity, which
limits its applications in some domains. Moreover, the mutual
coupling between the vertical and horizontal directions of the
element is neglected, resulting in additional phase errors.
Design of dual-CP reflectarrays with a dual-CP feed have
always been a challenge due to the complexity of the design
and fabrication of the dual-CP feeds. However, with a dual-
LP feed, the reflectarray can transform the dual-LP incidences
from the feed into dual-CP radiation, whose difficulty depends
on the phase relationship between the two orthogonal linear
polarizations. Although the operation of a dual-CP reflectarray
fed by a dual-LP feed was mentioned in [12], no dual-CP
reflectarray has been designed and fabricated to the best of
our knowledge.
0018-926X © 2016 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.
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2. 4236 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 10, OCTOBER 2016
Fig. 1. Illustration of LP–CP transformation. (a) Incident LP waves.
(b) Reflected CP waves.
II. EFFECTS OFMUTUALCOUPLING ONREFLECTARRAY
ELEMENTS FORCP RADIATION
The mechanism of LP to CP transformation is described in
Fig. 1. The polarization of an LP feed is along the diagonal
direction of the unit cell in a reflectarray aperture. As the LP
incidence waves from the feed impinge onto the reflectarray,
it is decomposed into two componentsE
in
x
andE
in
y
with equal
amplitude as shown in Fig. 1(a). In the case of lossless and
ground-backed structure, the incident waves are completely
reflected so that the reflected waves also have equal amplitude
with|E
out
x
|=|E
out
y
|. After achieving 90° phase difference
between the two orthogonal vector components of the reflected
waves by adjusting thex-andy-axis dimensions of the
element, the LP incident wavesfrom the feed can be converted
into a CP collimated beam, as shown in Fig. 1(b). If the
polarization of the LP incident waves is along the other
diagonal direction of the unit cell,−90° phase difference
between the two orthogonal vector components of the reflected
waves will be achieved, corresponding to the other handedness
of CP collimated beam. Therefore, the dual-CP reflectarray can
be implemented using a dual-LP feed.
Different from element design for the LP reflectarray or
CP reflectarray with a CP feed, the mutual coupling between
the two orthogonal polarizations of the CP element should be
taken into account in this case. It is the interaction between
the two orthogonal polarizations of the element, depending
on the element structures, that makes the independent control
of two orthogonal polarizations difficult, which will also
result in lower efficiency and narrower AR bandwidth for
CP reflectarray. Fortunately, similar to frequency selective
surfaces, the reflectarray elements can be arranged into four
catalogs: center connected types, solid interior types, loop
types and combination types [13]. The mutual coupling of
the CP reflectarray elements for the former three types is
discussed to analyze the mechanism of mutual coupling in
this section. The interference between the two orthogonal field
components of the last type can also be inferred from the three
formers because it can be viewed as a combination of the three
fundamental structures.
A simple example of the center connected type element
is cross dipole which was used as the unit cell in [11] for
CP radiation. The coupling between the thin cross dipole
arms is negligible, but stronger for wider arms cross dipole,
which can be explained as follows. For thin cross dipole,
the currents are gradually concentrated onto the central part
of vertical dipole for verticalpolarization incidence, which
can be viewed as a short circuit point equivalent. Therefore,
the phase compensation in thevertical polarization can be
achieved by adjusting the length of the vertical dipole, making
the independent control of the two orthogonal polarizations
possible. However, as the dipole width increases, currents are
induced on the horizontal dipole for the vertical incidence
and the central part of vertical dipole cannot be viewed as
a short circuit point anymore so that the horizontal dipole
will interfere the vertical polarized reflection characteristics.
Generally, the mutual coupling between the two orthogonal
polarizations of the element will be weaker as the width of
the dipole arms decreases, however, the slope of the phase
response curve will increase simultaneously, leading to a
narrower element bandwidth.
One of the typical solid interior type elements is rectangular
patch that was used as the unit cell in [7]. The coupling is
strongest near the resonant frequency, with maximal phase
variations of about 80° as shown in [7]. The mutual coupling
between the two orthogonal polarizations of the rectangular
patch is generally stronger than cross dipole, because the
rectangular patch can be viewed as cross dipole if the width
of cross-dipole arms is equal to the orthogonal dimension of
the cross dipole.
The typical loop type element is rectangular loop. The
mutual coupling between the two orthogonal polarizations
of the loop element is stronger than the cross-dipole and
rectangular patch elements because the loop structure reasons
when its perimeter is close to one wavelength and thus the
vertical polarization reflection phase of the loop element
directly depends on the horizontal dimension.
III. P
ROPOSEDELEMENT
The most severe drawback of reflectarrays is their narrow
operating band which is limited primarily by two factors. The
first one is the intrinsically narrow bandwidth of the microstrip
elements. The second is the frequency dependence of the
phase delay of the incident field [1]. In particular, the first
factor is quite critical and becomes dominant in moderate size
reflectarrays. It has been shown recently that the bandwidth
of reflectarray can be improved by using subwavelength ele-
ments [14]–[16]. Loop structure is superior in subwavelength
reflectarray designs because ofits smallest electrical resonant
dimensions compared to cross-dipole and patch elements,
which permits a denser lattice for a reflectarray comprised of
loop elements. Moreover, various single-layer multiresonance
elements in literature are comprised of loop structure to avoid
physical interferences and attain an adequate phase range
exceeding 360° [17]–[20]. Unfortunately, mutual coupling and
interference between the two orthogonal polarizations for loop
type structure is strongest as discussed in Section II. Therefore,
the design process of subwavelength multiresonant CP element
becomes more complicated without the help of loop structure.
Configurations of the reflectarray element adopted in this
paper are shown in Fig. 2(a) and (b). The square unit cell
has a lattice sizeP=8.33 mm, corresponding toλ
0/3
(λ
0is the free-space wavelength at 12 GHz). F4BM substrate
(ε
r=2.2 and dielectric loss tangent tanζ=0.002) is used

3. WUet al.: BROADBAND, SINGLE-LAYER DUAL CP REFLECTARRAYS WITH LINEARLY POLARIZED FEED 4237
Fig. 2. Geometry of the subwavelength multiresonance element. (a) Top
view. (b) Side view.
and substrate thicknessh=3 mm. The proposed subwave-
length multiresonance element is a combination type structure
consisting of a Jerusalem cross and an open loop, whose
mutual coupling between the two orthogonal polarizations
can be analyzed from the three basic structure as introduced
in Section II.
First, the Jerusalem cross is a center connected type struc-
ture. A thin width of the Jerusalem cross armsw=0.2mm
is set to attain a short-circuit point equivalently at the central
part of the Jerusalem cross and to achieve low mutual coupling
between the two orthogonal field components for this type of
structure as introduced in Section II. Second, the Jerusalem
cross consists of a crossed dipole with end loading, and the
current in the vertical conductors flows to charge the capacitors
formed by the horizontal conductors. This maintains a lower
resonant frequency compared to cross-dipole and patch ele-
ments, leading to a possibility of a denser lattice, which means
that it can replace loop structure in subwavelength reflectarray
design and maintain weak mutual coupling between the two
orthogonal field components simultaneously. Third, another
advantage of the Jerusalem cross against the loop element
is it features more geometrical parameters, which can be
adjusted to attain broader element bandwidth. Therefore, the
ratioM=b
1/Lx,N=b 2/Lyare introduced to achieve a
smaller slope of the reflection phase curve.
The open loop is intentionally employed to change the
initial current distribution of the loop structure and transform
the loop type into a connected type structure, and thus the
cross coupling between the two orthogonal polarizations is
expected to be low because of the independent control of
the two orthogonal polarization. Moreover, the open loop
can also avoid physical interference as with loop structure
and resonates at high frequency simultaneously. Therefore,
the multiresonance of the propose element can provide an
adequate phase range excessing 360°.
The other parameters of the element have been optimized to
achieve a smaller slope of the phase reflection curve. The gap
between the Jerusalem cross and open loop g is set to 0.2 mm
to attain strong mutual coupling between them, which will
improve the linearity of the reflection phase curve. The final
design parameters are as follows:
a
1=Lx,a2=Ly,M=0.5,N=0.5.
Fig. 3. Reflection phase and currentdistribution of the proposed element.
Fig. 4. Reflection phase of the proposed element with plane wave incidence
at various angles at different frequency.
The elements are modelled in HFSS simulator [21]
with periodic boundary conditions. The reflected phase
of the normalx-polarized incidence versusL
xat 12
GHz and the current distribution on the metallic pattern
are shown in Fig. 3. It is seen that the maximal phase
variation is 20° approximately, indicating the low mutual
coupling between the two orthogonal polarizations of the
proposed element. The reflected phase versusx-directional
dimensionL
xwith a fixedy-directional dimension
L
y=2.8 mm for different angles and polarizations
(TM,ϕ=0° and TE,ϕ=90°) at different frequencies are
shown in Fig. 4. As it is shown, the reflection phase curves
are not significantly affected bythe incident angles, with the
maximal phase discrepancy of about 14° at frequency around
13 GHz forθ=30° incidence. In addition, only the edge
elements are illuminated at large angle, thus the reflectarray
was simply designed based on the reflection phase of normal
incident plane waves.
IV. D
UAL-CP REFLECTARRAYDESIGN
The phase matching method where the array elements match
the required phase at multiple frequencies is an efficient
approach for the design of broadband reflectarrays [22]–[24].
In this paper, the phase matching is applied on the orthogonal
polarizations to achieve a higher degree of circular polariza-
tion. A 20×20-element dual-CP reflectarray with a dual-LP
feed centered at 12 GHz has been designed as shown in Fig. 5.
A dual-LP quadri-ridged horn antenna is used as the feed, and
is oriented toward the geometrical center of the reflectarray.

4. 4238 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 10, OCTOBER 2016
Fig. 5. Geometry of the dual-CP reflectarray antenna.
To minimize the feed blockage, the horn is tilted by 30° and
the main beam direction is normal to the reflectarray aperture.
The focus-to-diameter ratio (F/D) of 0.84 is chosen to provide
a proper illumination with−10 dB edge taper. The location
of the feed with respect to the center of the reflectarray is as
follows:
X
f=−81 mm,Y f=−81 mm,andZ f=198 mm.
The required compensation phase for each element in the
reflectarray aperture is calculated by
ϕ
mn=k(R mn−

rmn·ˆr0)+ϕ (1)
whereR
mnis the spatial distance between the feed and the
mnth element,ˆr
0is the unit vector in the main beam direction,

rmnis the position vector of themnth element andϕis a
constant reference phase. The desired phase distributions of the
two orthogonal polarizations are calculated using formula (1)
and the calculated results are shown in Fig. 6. Based on
the reflected phase of thex-polarized incidence versusL
xat
12 GHz as shown in Fig. 3, a 2-D interpolation function can be
used to calculate the reflected phase of the proposed element
with different combination of (L
x,Ly). An objective function
about the compensation phase errors is defined
obj
mn
(Lx,Ly)=|f mn(Lx,Ly)−phase
mn
|
+
≤
≤
≤f
mn(Lx,Ly)−phase
mn
±
π
2
≤
≤
≤(2)
where obj
mn(Lx,Ly)is the sum of the differences between
the desired and the achievable phase of the two orthogonal
polarizations for the mnth element,f
mn(Lx,Ly)is the
achievable phase delays forx-polarization,f
mn(Ly,Lx)is
the achievable phase delays fory-polarization symmetrically
and phase
mnand phasemn±(π/2)represent the desired phase
delays forx-polarization andy-polarization, respectively, as
shown in Fig. 6. Then, the process of matching the achievable
to the desired phase can be turned into an optimization subject
to bounds on variables
min obj
mn
(Lx,Ly)
s.t.L
xmin
≤Lx≤Lxmax
,Lymin
≤Ly≤Lymax
.
Finally, the most suitable parameters (L
x,Ly) can be found
out to minimize the phase errors function by a simple program.
Fig. 6. Desired phase distributions along (a)x-direction and (b)y-direction.
Fig. 7. Minimal phase error for dual-CP reflectarray with subwavelength
multiresonance elements.
Fig. 8. Photograph of the dual-CP reflectarray using subwavelength
multiresonance elements.
The minimal phase errors function objmn(Lx,Ly) is illustrated
in Fig. 7, with the maximal phase errors of about 5°.
V. S
IMULATED AND MEASUREDRESULTS
A single-layer dual-CP reflectarray with 20×20 opti-
mized subwavelength multiresonance elements introduced in
Section III has been designed, simulated and fabricated. The
designed reflectarray antenna is verified using Ansys HFSS
and the reflectarray phasing elements, along with the dual-
LP feed are modeled, as shown in Fig. 5. Photograph of
the fabricated antenna prototype is shown in Fig. 8. Note

5. WUet al.: BROADBAND, SINGLE-LAYER DUAL CP REFLECTARRAYS WITH LINEARLY POLARIZED FEED 4239
Fig. 9. Measured and simulated radiation patterns in the azimuth plane. Horn with (a) vertical and (b) horizontal polarizations at 10 GHz. Horn
with (c) vertical and (d) horizontal polarizations at 12 GHz. Horn with (e) vertical and (f) horizontal polarizations at 14 GHz.
that due to lack of dual-LP feed horn when measured in the
anechoic chamber, a single LP pyramidal horn is employed
for simplification and the orthogonal polarization is achieved
by rotating the horn 90°. During the measures, all fixtures are
covered by microwave absorbers except the reflectarray itself
and the feed horn. The measured and simulated normalized
radiation patterns of the reflectarray for both co-polarization
(co-pol) and cross-polarization (cr-pol) in the azimuth plane
at 10, 12 and 14 GHz are presented in Fig. 9. The measured
and simulated results show that the AR of the reflectarray
can maintain approximately a constant on the range of the
main lobe width in this case, and thus the AR in the broad-
side direction is employed to illustrate the level of cr-pol.
At 12 GHz, the measured cr-pol level is−33 dB below the
peak co-pol gain in the broadside direction, corresponding
to the AR of 0.38 dB and the sidelobe levels are 27 dB
down from the main beam at the center frequency as shown
in Fig. 9(c). By rotating the pyramidal horn with horizontal
polarization by 90°, the other handedness of CP radiation
with co-pol (RHCP) and cr-pol (LHCP) can be achieved
as shown in Fig. 9(d). The reflectarray also shows good
scattering results at other frequencies, which have not been
shown in Fig. 9 to make these figures more legible. The
measured and simulated gain and AR against the frequency
in the broadside direction are presented in Figs. 10 and 11,
respectively. In the fabrication process, the installation error
of the prototype and measurement error are the main reasons
of discrepancies between the measured and simulated results.
As shown in Fig. 10, the designed reflectarray can realize dual-
CP radiation, and a measured 1-dB gain bandwidth of 12.5%
(from 11.5 to 13 GHz) is achieved. The maximum gain is
24.4 dBi at 12.5 GHz, corresponding to the antenna aperture

6. 4240 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 64, NO. 10, OCTOBER 2016
Fig. 10. Measured and simulated gain for reflectarray using subwavelength
multiresonance elements.
Fig. 11. Measured and simulated ARs for the reflectarray using subwave-
length multiresonance elements. Horn with (a) vertical and (b) horizontal
polarizations.
efficiency of 46.3%. The 3-dB AR bandwidths of both dual
CP radiations are about 50% (from 9 to 15 GHz) as shown
in Fig. 11.
To compare the results, a dual-CP reflectarray composed
of cross-dipole elements with a fixed width of cross-dipole
armsw=3 mm was also designed and simulated. The
design frequency, aperture size, F/D, and the feed for this
reflectarray are the same as the reflectarray with the proposed
elements. The gain and AR of the dual-CP reflectarray in
the broadside direction against the frequency are shown in
Fig. 12. The simulated 1-dB gain bandwidth is about 8% (from
11.4 to 12.5 GHz) and the 3-dB AR bandwidth is about 9%
(from 11.2 to 12.3 GHz). Therefore, the bandwidths of the gain
and AR of the dual-CP reflectarray are improved significantly
Fig. 12. Simulated gain and AR of the dual-CP reflectarray using cross-dipole
elements.
by using subwavelength multiresonance elements compared to
cross-dipoles elements.
VI. C
ONCLUSION
A broadband single-layer dual-CP reflectarray with a dual-
LP feed was introduced in this paper. Based on an investi-
gation into the mutual coupling between the two orthogo-
nal polarizations of three fundamental types of reflectarray
elements, a novel subwavelength multiresonance element is
proposed which shows a linear phase response and provides
an adequate phase range over 360°. Also, the cross coupling
between the vertical and horizontal polarizations of the CP
element is minimized by an optimization technique. Measure-
ments of the fabricated 20×20-element reflectarray achieved
a 1-dB gain bandwidth of 12.5% and 3-dB AR bandwidth of
about 50%, which shows a significant improvement in gain
and AR bandwidth performance of the dual-CP reflectarray.
Additionally, the proposed subwavelength multiresonance ele-
ment and optimization method in this paper can also be used in
other applications, e.g., polarization separator and polarization
transformation in folded reflectarray.
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Geng-Bo Wuwas born in Guangdong, China, in
1992. He received the B.Eng. degree in electro-
magnetics wave propagation and antenna from the
University of Electronic Science and Technology of
China, Chengdu, China, in 2015, where he is cur-
rently pursuing the M.Sc. degree in electromagnetics
and microwave technology.
His current research interests include reflectarray
antenna and phased arrays.
Shi-Wei Qu(S’08–M’11–SM’12) was born in
He’nan, China, in 1980. He received the B.Eng.
and M.Sc. degrees from the University of Elec-
tronic Science and Technology of China (UESTC),
Chengdu, China, in 2001 and 2006, respectively,
and the Ph.D. degree from the City University of
Hong Kong (CityU), Hong Kong, in 2009.
He was with the 10th Institute of Chinese Informa-
tion Industry from 2001 to 2002. He was a Research
Assistant with the Department of Electronic Engi-
neering, CityU, from 2006 to 2007. He was a COE
(Global Center of Excellence) Research Fellow and a Post-Doctoral Fellow
with Tohoku University, Sendai, Japan, from 2009 to 2010. He is currently
an External Member with the State Key Laboratory of Millimeter Waves,
Partner Laboratory, CityU. He is a Full Professor with the School of Electronic
Engineering, UESTC. He has authoredor co-authored over 60 internationally
refereed papers and over 30 international conference papers. His current
research interests include UWB antennas and arrays, phased arrays, and
millimeter-wave/terahertz antennas and arrays.
Shiwen Yang (M’00–SM’04) received the
B.Sc. degree in electronic science from East
China Normal University, Shanghai, China, in
1989, and the M.Eng. degree in electromagnetics
and microwave technology and the Ph.D. degree
in physical electronics from the University of
Electronic Science and Technology of China
(UESTC), Chengdu, China, in 1992 and 1998,
respectively.
He was a Lecturer with the Institute of
High Energy Electronics, UESTC, from
1994 to 1998. He was a Research Fellow with the School of Electrical
and Electronic Engineering, Nanyang Technological University, Singapore,
from 1998 to 2001. He joined Temasek Laboratories, National University
of Singapore, Singapore, as a Research Scientist in 2002. He is currently
a Full Professor with the Department of Microwave Engineering, School
of Electronic Engineering, UESTC. His current research interests include
antennas, antennas arrays, and computational electromagnetics.
Chi Hou Chan(S’86–M’86–SM’00–F’02) received
the B.S. and M.S. degrees from The Ohio State
University, Columbus, OH, USA, in 1981 and 1982,
respectively, and the Ph.D. degree from the Univer-
sity of Illinois at Urbana–Champaign, Urbana, IL,
USA, in 1987, all in electrical engineering.
He was a Visiting Assistant Professor with the
Department of Electrical and Computer Engineering,
University of Illinois at Urbana–Champaign, from
1987 to 1989. He was a Faculty Member with
the Department of Electrical Engineering, University
of Washington, Seattle, WA, USA, from 1989 to 1998. He joined the
Department of Electronic Engineering,City University of Hong Kong (CityU),
Hong Kong, in 1996, where he was promoted to Chair Professor of Electronic
Engineering in 1998. He was first an Associate Dean and then the Dean of
the College of Science and Engineering from 1998 to 2009. He served as an
Acting Provost with CityU from 2009 to 2010. He is currently the Director
of the State Key Laboratory of Millimeter Waves with the Partner Laboratory,
CityU. His current research interests include computational electromagnetics,
millimeter-wave circuits and antennas,and terahertz science and technology.
Dr. Chan received the U.S. National Science Foundation Presidential Young
Investigator Award in 1991, and the Joint Research Fund for Hong Kong
and Macao Young Scholars and the National Science Fund for Distinguished
Young Scholars, China, in 2004. He received outstanding teacher awards from
the Department of Electronic Engineering, CityU, in 1998, 1999, 2000, and
2008. He is the General Co-Chair of ISAP 2010, iWAT2011, iWEN 2013,
ICCEM 2015, and ICCEM 2016.